Sampled amplitude read channel employing pipelined reads to reduce the gap between sectors

ABSTRACT

A sampled amplitude read channel incorporated within a magnetic disk storage system for reading data recorded tracks on a magnetic medium, where the data comprises user data sectors recorded at varying data rates across a plurality of predefined zones and embedded servo data sectors recorded at the same data rate across the zones. The read channel comprises a timing recovery component for synchronous sampling of a read signal from a magnetic read head positioned over the magnetic medium, a gain control component for adjusting the amplitude of the read signal, and a DC offset component for canceling a DC offset in the read signal. These components are dynamically configured to operate according to whether the read channel is processing user data or embedded servo data.

This application is a divisional of application Ser. No. 08/440,515filed on May 12, 1995 now U.S. Pat. No. 5,796,535.

FIELD OF INVENTION

The present invention relates to the control of magnetic storage systemsfor digital computers, and particularly, to discrete time circuitryintegrated into a sampled amplitude read channel for synchronousdetection of user data and embedded servo data.

CROSS REFERENCE TO RELATED APPLICATIONS AND PATENTS

This application is related to several U.S. patents, namely U.S. Pat.No. 5,424,881 entitled “Synchronous Read Channel,” U.S. Pat. No.5,359,631 entitled “Timing Recovery Circuit for Synchronous WaveformSampling,” U.S. Pat. No. 5,291,499 entitled “Method and Apparatus forReduced-Complexity Viterbi-Type Sequence Detectors,” U.S. Pat. No.5,297,184 entitled “Gain Control Circuit for Synchronous WaveformSampling,” and U.S. Pat. No. 5,329,554 entitled “Digital PulseDetector.” All of the above-named patents are assigned to the sameentity, and all are incorporated herein by reference.

BACKGROUND OF THE INVENTION

In magnetic disk storage systems for computers, digital data serves tomodulate the current in a read/write head coil so that a sequence ofcorresponding magnetic flux transitions are written onto a magneticmedium in concentric tracks. To read this recorded data, the read/writehead passes over the magnetic medium and transduces the magnetictransitions into pulses in an analog signal that alternate in polarity.These pulses are then decoded by read channel circuitry to reproduce thedigital data.

Decoding the pulses into a digital sequence can be performed by a simplepeak detector in a conventional analog read channel or, as in morerecent designs, by a discrete time sequence detector in a sampledamplitude read channel. Discrete time sequence detectors are preferredover simple analog pulse detectors because they compensate forintersymbol interference (ISI) and, are less susceptible to noise. As aresult, discrete time sequence detectors increase the capacity andreliability of the storage system.

There are several well known discrete time sequence detection methodsincluding discrete time pulse detection (DPD), partial response (PR)with Viterbi detection, maximum likelihood sequence detection (MLSD),decision-feedback equalization (DFE), enhanced decision-feedbackequalization (EDFE), and fixed-delay tree-search with decision-feedback(FDTS/DF).

In conventional peak detection schemes, analog circuitry, responsive tothreshold crossing or derivative information, detects peaks in thecontinuous time analog signal generated by the read head. The analogread signal is “segmented” into bit cell periods and interpreted duringthese segments of time. The presence of a peak during the bit cellperiod is detected as a “1” bit, whereas the absence of a peak isdetected as a “0” bit. The most common errors in detection occur whenthe bit cells are not correctly aligned with the analog pulse data.Timing recovery, then, adjusts the bit cell periods so that the peaksoccur in the center of the bit cells on average in order to minimizedetection errors. Since timing information is derived only when peaksare detected, the input data stream is normally run length limited (RLL)to limit the number of consecutive “0” bits.

As the pulses are packed closer together on the concentric data tracksin the effort to increase data density, detection errors can also occurdue to intersymbol interference, a distortion in the read signal causedby closely spaced overlapping pulses. This interference can cause a peakto shift out of its bit cell, or its magnitude to decrease, resulting ina detection error. The ISI effect is reduced by decreasing the datadensity or by employing an encoding scheme to ensure that a minimumnumber of “0” bits occur between “1” bits. For example, a (d,k) runlength limited (RLL) code constrains to d the minimum number of “0” bitsbetween “1” bits, and to k the maximum number of consecutive “0” bits. Atypical RLL code is a (1,7) ⅔ rate code which encodes 8 bit data wordsinto 12 bit codewords to satisfy the (1,7) constraint.

Sampled amplitude detection, such as partial response (PR) with Viterbidetection, allows for increased data density by compensating forintersymbol interference and increasing channel noise immunity. Unlikeconventional peak detection systems, sampled amplitude recording detectsdigital data by interpreting, at discrete time instances, the actualvalue of the pulse data. The analog pulses are sampled at the baud rate(code bit rate) and the digital data is detected from these discretetime sample values. A discrete time sequence detector, such as a Viterbidetector, interprets the discrete time sample values in context todetermine a most likely sequence for the data. In this manner, theeffect of ISI can be taken into account during the detection process,thereby decreasing the probability of a detection error. This increasesthe effective signal to noise ratio and, for a given (d,k) constraint,allows for significantly higher data density as compared to conventionalanalog peak detection read channels.

The application of sampled amplitude techniques to digital communicationchannels is well documented. See Y. Kabal and S. Pasupathy, “PartialResponse Signaling”, IEEE Trans. Commun. Tech., Vol. COM-23, pp.921-934,September 1975; and Edward A. Lee and David G. Messerschmitt, “DigitalCommunication”, Kluwer Academic Publishers, Boston, 1990; and G. D.Forney, Jr., “The Viterbi Algorithm”, Proc. IEEE, Vol. 61, pp. 268-278,March 1973.

Applying sampled amplitude techniques to magnetic storage systems isalso well documented. See Roy D. Cideciyan, Francois Dolivo, WalterHirt, and Wolfgang Schott, “A PRML System for Digital MagneticRecording”, IEEE Journal on Selected Areas in Communications, Vol. 10No. 1, January 1992, pp.38-56; and Wood et al, “Viterbi Detection ofClass IV Partial Response on a Magnetic Recording Channel”, IEEE Trans.Commun., Vol. Com-34, No. 5, pp. 454-461, May 1986; and Coker Et al,“Implementation of PRML in a Rigid Disk Drive”, IEEE Trans. onMagnetics, Vol. 27, No. 6, November 1991; and Carley et al, “AdaptiveContinous-Time Equalization Followed By FDTS/DF Sequence Detection”,Digest of The Magnetic Recording Conference, Aug. 15-17, 1994, pp. C3;and Moon et al, “Constrained-Complexity Equalizer Design for Fixed DelayTree Search with Decision Feedback”, IEEE Trans. on Magnetics, Vol. 30,No. 5, September 1994; and Abbott et al, “Timing Recovery For AdaptiveDecision Feedback Equalization of The Magnetic Storage Channel”,Globecom'90 IEEE Global Telecommunications Conference 1990, San Diego,Calif., November 1990, pp.1794-1799; and Abbott et al, “Performance ofDigital Magnetic Recording with Equalization and Offtrack Interference”,IEEE Transactions on Magnetics, Vol. 27, No. 1, January 1991; and Cioffiet al, “Adaptive Equalization in Magnetic-Disk Storage Channels”, IEEECommunication Magazine, February 1990; and Roger Wood, “EnhancedDecision Feedback Equalization”, Intermag'90.

Similar to conventional peak detection systems, sampled amplitudedetection requires timing recovery in order to correctly extract thedigital sequence. Rather than process the continuous signal to alignpeaks to the center of bit cell periods, as in peak detection systems,sampled amplitude systems synchronize the sampling of the pulses to thebaud rate. That is, timing recovery adjusts the sampling clock in orderto minimize the error between the signal sample values and estimatedsample values. A pulse detector or slicer determines the estimatedsample values from the read signal samples. Even in the presence of ISIthe sample values can be estimated and, together with the signal samplevalues, used to synchronize the sampling of the analog pulses in adecision-directed feedback system.

The decision-directed feedback system is normally implemented using aphase-locked-loop (PLL) circuit comprising a phase detector forgenerating a phase error based on the difference between the estimatedsamples and the read signal samples. A loop filter filters the phaseerror, and the filtered phase error operates to adjust the samplingclock which is typically the output of a variable frequency oscillator(VFO) with the filtered phase error as the control input. The output ofthe VFO controls the sampling clock of a sampling device such as ananalog-to-digital (A/D) converter.

It is helpful to first lock the PLL to a reference or nominal samplingfrequency so that the desired sampling frequency, with respect to theanalog pulses representing the digital data, can be acquired and trackedmore efficiently. The nominal sampling frequency is the baud rate, therate that data was written onto the medium. Therefore, one method tolock-to-reference is to generate a sinusoidal signal relative to thewrite clock and inject this signal into the PLL. Once locked to thereference frequency (write frequency), the PLL input switches from thewrite clock to the signal from the read head in order to synchronize thesampling of the waveform in response to a sinusoidal acquisitionpreamble recorded on the medium.

The timing recovery loop filter controls the dynamics of thephase-lock-loop. Accordingly, the loop filter coefficients are adjustedto achieve a desired transient response and tracking quality. For goodtracking quality, the loop bandwidth should be narrow so that phasenoise and gain variance are attenuated. During acquisition, the loopbandwidth should be as wide as possible without being unstable toachieve a fast transient response in order to minimize the length of theacquisition preamble.

Sampled amplitude read channels also employ a decision-directed feedbacksystem to control the gain of the analog read signal in order tominimize a gain error between the signal sample values and estimatedsample values. The gain error is filtered and applied to the controlinput of a variable gain amplifier (VGA) in order to adjust themagnitude of the read signal toward the desired partial response.

A DC offset in the analog read signal can adversely affect theperformance of the automatic gain control and timing recovery circuitry.A decimation DC offset circuit can compensate for the undesirableeffects by computing the DC offset from the read signal sample valuesand subtracting it from the analog read signal before sampling (see,e.g., the above referenced co-pending U.S. patent application Ser. No.08/341,251). Similar to the gain control and timing recovery loops, theDC offset loop comprises a filter for filtering the computed DC offset.

Also relevant to the present invention is the servo control system forpositioning the read/write head over a selected track in order to readand write information. In disk drives utilizing either analog or sampledamplitude read channels, the read/write head is normally mounted on anactuator arm which is positioned by means of a voice coil motor (VCM).The VCM moves the head and actuator arm assembly across the disk surfaceat a very high speed to perform seek operations in which the head ispositioned over a selected data track. The VCM also maintains the headover the selected track while reading or writing information assuccessive portions of the track pass under the head. A servo systemcontroller provides the head positioning necessary for reading andwriting information in response to requests from a computer to which thedisk drive is connected.

In embedded servo disk drives, servo fields are normally recorded on thedisk as radial spokes 17 that “split” the data sectors 15 as shown inFIG. 2A. Each servo spoke 17 is referred to as a “wedge” of servo datacomprising servo control information and servo data. The servo controlinformation typically includes a preamble 5 to allow gain control toacquire to the read signal before reading the servo data, and a servosynch mark 7 to signal the beginning of the servo data 3 (see FIG. 2B).If the servo data is detected synchronously, then the preamble is alsoused to synchronize timing recovery to the read signal before readingthe servo data. The servo data may also comprise a track number codewhich is a Gray coded integer value of the track currently spanned bythe read/write head, a head number identifying the current platter in amulti-disk system, and a wedge number identifying the current servowedge. The servo data may also optionally comprise a servo address markfor asynchronous identification of the wedge when the disk drive spinsup initially.

The embedded servo field also typically includes off-track burstinformation physically positioned at precise intervals and locationswith respect to the various track centerlines to provide the servosystem controller with information relative to the fractionaltrack-to-track displacement of the head with respect to a selected trackcenterline. The servo controller uses the servo burst information tokeep the read head aligned over the centerline of the selected trackwhile data is written or read from the medium.

Similar to the servo data sectors, the user data sectors 15 alsocomprise an acquisition preamble 68 and a sync mark 70 to signal thebeginning of a user data field 72 as shown in FIG. 2B.

Zoned recording is a technique known in the art for increasing thestorage density by recording the user data at different rates inpredefined zones between the inner diameter (ID) and outer diameter (OD)tracks. The data rate can be increased at the outer diameter tracks dueto the increase in circumferential recording area and the decrease inintersymbol interference. This allows more data to be stored in theouter diameter tracks as is illustrated in FIG. 2A where the disk ispartitioned into an outer zone 11 comprising fourteen data sectors pertrack, and an inner zone 13 comprising seven data sectors per track. Inpractice, the disk may actually be partitioned into several zones atvarying data rates.

When using synchronous detection to read the servo fields, the readchannel operates in the same manner as if it were reading user data asdescribed above. That is, the gain and DC offset circuitry adjust theamplitude and offset of the analog read signal, and timing recoverylocks to a reference frequency, acquires the servo preamble, andsynchronizes the sampling of the servo data. In most disk drive storagesystems, however, the embedded servo fields are not recorded at the samerate as the user data. There is, therefore, a deficiency in theoperation of read channels that cannot re-program the gain control,timing recovery, and DC offset circuits when the read channeltransitions between reading user data and servo data.

Other drawbacks overcome by the present invention include: the inabilityto pipeline reads to reduce the gap between sectors; the inability touse the read channel sampling device for sampling other analog signalsgenerated by the storage system such as servo control signals; theinability to use information provided in the preamble field to optimizeoperation of the sync mark detector; the inability to detect the userdata sync mark and the servo data sync mark using the same sync markdetector; and the inability to adjust the response of the gain controlloop when searching for the servo address mark.

What is needed is a robust technique for re-programming the gaincontrol, DC offset control, and timing recovery control circuits whenthe read channel transitions between reading user data and embeddedservo data. Another object is to pipeline reads to minimize the gap onthe medium between adjacent user data sectors and the gap between userdata and servo data sectors. A further object is to use the read channelsampling device for sampling auxiliary analog signals generated by thestorage system. Another object is to use the same sync mark detector fordetecting the user data sync mark and the servo data sync mark. Yetanother object is to enable operation of a sync mark detector at a clockinterval selected in relation to the preamble. A further object is toadjust the frequency response of the gain control loop when searchingfor the servo address mark.

SUMMARY OF THE INVENTION

In a sampled amplitude read channel for reading user data and embeddedservo data in a magnetic disk storage device, a plurality of componentsincluding a timing recovery circuit, an automatic gain control circuit,and a DC offset circuit comprise at least one filter programmed from aset of “shadow” registers corresponding to whether the read channel isin a user data or servo data mode. The filter coefficients andaccumulation paths are updated from the shadow registers when the readchannel transitions between reading user data and embedded servo data.The magnetic disk is partitioned into several zones, and the filters areinitialized with calibrated values when the read/write head passes intoa new zone.

The timing recovery circuit comprises a phase-locked-loop forsynchronizing the sampling of an analog read signal generated from theread head passing over the magnetic medium. The timing recovery PLLincludes a variable frequency oscillator (VFO) comprising an output forcontrolling the sampling frequency of a sampling device; a first controlinput for receiving a channel data rate (CDR) command; and a secondcontrol input for receiving a center frequency command. The CDR commandis programmably set from a user/servo shadow register according to thezone where the selected track is located.

The PLL center frequency command comprises a course setting, a biassetting and a fine setting. The course setting is generated either by auser data synthesizer or a servo data synthesizer depending on theuser/servo mode of the read channel. The bias setting is programmablyset from a user/servo shadow register and compensates for differences infabrication between the synthesizer VFOs and the timing recovery VFO.The fine setting is generated by a discrete time phase error detectorwhich measures a difference between the sampling phase/frequency and thebaud rate.

Before acquiring the acquisition preamble preceding the user or servodata, the timing recovery PLL is locked to the output of the user orservo synthesizer depending on the user/servo mode. This is accomplishedby injecting, into the read channel, the analog output signal from therespective synthesizer so that timing recovery locks onto theappropriate frequency.

In order to reduce the gap between adjacent user data sectors and thegap between user data and servo data sectors, operation of the readchannel is pipelined by resetting the gain, timing recovery, and DCoffset circuits before the discrete time equalizing filter and sequencedetector have finished processing the samples for the current sector.This allows the read channel to begin acquiring the preamble of a nextsector (user or servo data) concurrent with processing the end of theprevious sector, thereby decreasing the physical gap on the mediumbetween sectors.

The read channel also comprises a programmable sync mark detector fordetecting both the user data and servo data sync marks in order to framethe operation of respective RLL user data and servo data decoders. Thesync mark detector is enabled by the timing recovery circuit at a sampleinterval predetermined in relation to the preamble field.

When the disk drive initially spins up, an asynchronous servo addressmark detector determines the location of the servo wedges with respectto the read head. To facilitate asynchronous detection, the gain controlcircuit computes a gain error according to a predetermined set point andthe maximum absolute value over a programmable block length.

The above and other advantages of the present invention will be betterunderstood with reference to the accompanying drawings together with thefollowing detailed description of the preferred embodiments.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a conventional sampled amplitude recordingchannel.

FIG. 2A shows an exemplary data format of a magnetic disk having aplurality of concentric tracks recorded in zones at varying data rateswhere each track contains a plurality of user data and embedded servodata sectors.

FIG. 2B shows an exemplary format of a user data sector and an embeddedservo data sector.

FIG. 3 is a block diagram of the sampled amplitude read channel of thepresent invention comprising automatic gain control, DC offset control,timing recovery, a first and second synthesizer for processing user andservo data respectively, an asynchronous servo address mark detector,and a sync mark detector for detecting user data and servo data syncmarks.

FIG. 4 shows an implementation of the user data and servo datasynthesizers.

FIG. 5 shows the DC offset control circuit comprising a filter having aprogrammable gain and accumulation path which are updated when the readchannel transitions between reading user data and servo data.

FIG. 6 shows the gain control circuit comprising a filter having aprogrammable gain and accumulation path which are updated when the readchannel transitions between reading user data and servo data.

FIG. 7 is a block diagram of the timing recovery circuit comprising aVFO for controlling the sampling frequency wherein the VFO is responsiveto a zero phase start signal and a frequency control signal generated bysumming a course center frequency setting, a fine center frequencysetting and a bias setting.

FIG. 8 is a detailed diagram of a preferred embodiment for the timingrecovery PID filter where the filter's coefficients and accumulationpath are updated between user and servo data modes.

FIG. 9 shows more details of the data/servo sync detector andparticularly the operation with respect to the timing recovery controlsignal.

FIG. 10A shows the acquisition read signal with corresponding actual andestimated sample values.

FIG. 10B is a detailed diagram of the preferred embodiment for theexpected sample value generator and phase error detector used in thetiming recovery circuit.

FIG. 11 is an alternative embodiment for the data/servo sync detectorwhich processes the even and odd interleaves of the read signal inparallel.

FIG. 12 is a simplified and enlarged illustration of a read/write head(shown in phantom) along a track centerline prior to the off-track servoburst passing underneath the head due to the rotation of the disk.

FIG. 13 illustrates a conventional analog read channel and associatedservo controller, the former incorporating an analog pulse detector andanalog servo burst measurement circuit for providing digital servo dataand analog burst signals, respectively, to the servo controller.

FIG. 14 illustrates a sampled amplitude read channel and associatedservo controller in accordance with the discrete time servo demodulatorcircuit of the present invention wherein the read channel comprises adiscrete time area detection circuit and discrete time pulse detectioncircuit for providing the servo controller with digital informationrepresentative of the servo burst and servo data, respectively, andwherein the servo controller need not incorporate an additional ADCcircuit.

FIG. 15 is a functional block diagram of a peak detector circuit for usein conjunction with the discrete time servo demodulator of the presentinvention.

FIG. 16 is a more detailed logic block diagram of the peak detectoroperable to produce output data in one of four modes of operation.

FIG. 17 illustrates the operation of the peak detector when processingthe sampled analog signal from the magnetic read head.

FIG. 18 is a simplified logic block diagram of a frequency dithercircuit which allows the sampling frequency to be changed over a set offrequencies within a small fraction of a nominal sampling rate.

FIGS. 19A-19D illustrate a sampled servo burst signal and correspondingrectified burst samples, squared burst samples and interpolated burstsamples respectively, useful for understanding the principles of thediscrete time area detect circuit of the discrete time servo demodulatorcircuit of the present invention.

FIG. 20 is a simplified logic block diagram of a discrete time areadetect circuit wherein the signal samples are first passed through anon-linearity block and then summed in an accumulator prior to beingstored in a corresponding register.

FIG. 21 is a logic block diagram of a portion of the discrete time areacircuit of the present invention wherein all samples and interpolatedsamples are rectified and accumulated to produce a servo burst amplitudemeasurement with the signal samples being processed in parallel.

FIG. 22 is a simplified logic block diagram of a servo burst accumulatorin accordance with the present invention for use in conjunction with thediscrete time area detection circuit.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT Conventional SampledAmplitude Read Channel

FIG. 1 is a detailed block diagram of a conventional sampled amplituderead channel. During a write operation, either user data 2 or preambledata from a data generator 4 (for example 2T preamble data) is writtenonto the media. An RLL encoder 6 encodes the user data 2 into a binarysequence b(n) 8 according to an RLL constraint. A precoder 10 precodesthe binary sequence b(n) 8 in order to compensate for the transferfunction of the recording channel 18 and equalizing filters to form aprecoded sequence ˜b(n) 12. The precoded sequence ˜b(n) 12 is convertedinto symbols a(n) 16 by translating 14 ˜b(N)=0 into a(N)=−1, and ˜b(N)=1into a(N)=+1. Write circuitry 9 responsive to the symbols a(n) 16,modulates the current in the recording head coil at the baud rate 1/T torecord the binary sequence onto the media. A frequency synthesizer 52provides a baud rate write clock 54 to the write circuitry 9.

When reading the recorded binary sequence from the media, timingrecovery 28 first locks to the write frequency by selecting, as theinput to the read channel, the write clock 54 through a multiplexor 60.Once locked to the write frequency, the multiplexor 60 selects thesignal 19 from the read head as the input to the read channel in orderto acquire the acquisition preamble. A variable gain amplifier 22adjusts the amplitude of the analog read signal 58, and an analog filter20 provides initial equalization toward the desired response. A samplingdevice 24 samples the analog read signal 62 from the analog filter 20,and a discrete time filter 26 provides further equalization of thesample values 25 toward the desired response. In partial responserecording, for example, the desired response is often selected fromTable 1. A DC offset circuit 1 responsive to the equalized sample values32 computes and subtracts the DC offset 29 from the analog read signal62.

The equalized sample values 32 are applied to decision directed gaincontrol 50 and timing recovery 28 for adjusting the amplitude of theread signal 58 and the frequency and phase of the sampling device 24,respectively. Timing recovery adjusts the frequency of sampling device24 over line 23 in order to synchronize the equalized samples 32 to thebaud rate. Frequency synthesizer 52 provides a coarse center frequencysetting to the timing recovery circuit 28 over line 64 in order tocenter the timing recovery frequency over temperature, voltage, andprocess variations. Gain control 50 adjusts the gain of variable gainamplifier 22 over line 21. The equalized samples Y(n) 32 are sent to adiscrete time sequence detector 34, such as a maximum likelihood (ML)Viterbi sequence detector, to detect an estimated binary sequence{circumflex over ( )}b(n) 33. An RLL decoder 36 decodes the estimatedbinary sequence {circumflex over ( )}b(n) 33 into estimated user data37. A data sync mark detector 66 detects the sync mark 70 (shown in FIG.2B) in the data sector 15 in order to frame the operation of the RLLdecoder 36 and signal the beginning of user data 72. In the absence oferrors, the estimated binary sequence {circumflex over ( )}b(n) 33 isequal to the recorded binary sequence b(n) 8, and the decoded user data37 is equal to the recorded user data 2.

Improved Sampled Amplitude Read Channel

FIG. 3 is a block diagram of the improved sampled amplitude read channelof the present invention comprised of a user data frequency synthesizerA100 and a servo data frequency synthesizer A102. When reading userdata, a control line U/S selects the output A114 of the user datasynthesizer A100 as the lock to reference frequency through amultiplexor A104. The control line U/S also selects the coarse centerfrequency setting A110 through a multiplexor A112 as the timing recoverycontrol signal 64. When the read channel switches into servo data modein order to read a servo wedge, the control line U/S selects the outputA106 of the servo data synthesizer A102 as the lock to referencefrequency through multiplexor A104. The control line U/S also selectsthe coarse center frequency setting A108 from the servo data synthesizerA102 through multiplexor A112 as the timing recovery control signal 64.

The read channel further comprises an asynchronous servo address markdetector A126 for generating a control signal A118 indicating when theservo address mark has been detected. The servo address mark detectorA126 switches operation of the gain control circuit over line A118 tocompensate for the unpredictable amplitude fluctuations caused by theinter-track head position and the wide range of user to servo datadensities.

A data/servo sync detector A120, responsive to the detected binarysequence 33 from the sequence detector 34, detects both user data andservo data sync marks and generates framing signals (A121,A119) to frameoperation of a user data RLL decoder 36 and a servo data RLL decoderA122, respectively. The sync detector A120 is also responsive to acontrol signal A124 from the timing recovery circuit 28 to aid in thesync detection process.

The read channel further comprises auxiliary analog inputs for samplingother analog signals generated within the disk drive such as the drivingcurrent for a Voice Coil Motor in a servo system, or the output of atemperature sensor. A multiplexor A101 selects, as the input to samplingdevice 24, the analog read signal 62 from the analog receive filter 20or one of a plurality of auxiliary input signals A103. When an auxiliaryinput is selected for sampling, the output 25 of the sampling device 24is stored into registers for subsequent processing by a microcontrollersuch as a servo controller.

Pipelined Reads

Before the read channel can process a new user or servo data sector,components such as the filters in timing recovery, gain control, and DCoffset control must be reconfigured to acquire the preamble of the newsector. During this reconfiguration process, the magnetic disk continuesto spin under the read head creating a physical gap on the mediumbetween the end of a current sector and the beginning of a new sector.In order to reduce the gap between sectors, operation of the readchannel is pipelined by reconfiguring the gain control 50, timingrecovery 28, and DC offset 1 circuits before the discrete timeequalizing filter 26 and sequence detector 34 have finished processingthe samples for the current sector. This allows the read channel tobegin acquiring the preamble (68,5) of a next sector (user or servodata) concurrent with processing the end of the previous sector, therebydecreasing the physical gap on the medium between sectors.

Dual Synthesizers

FIG. 4 shows a block diagram of the user data synthesizer A100 and theservo data synthesizer A102. Each synthesizer comprises a VFO (B130a,B130 b) that is part of a phase locked loop having a referencefrequency input B126 divided by a DIV Nx circuit (B128 a,B128 b). Phasedetectors (B132 a,B132 b) generate a respective phase error between theoutput of the DIV Nx circuits (B128 a,B128 b) and the output of the VFOs(B130 a,B130 b) divided by a DIV Mx circuit (B138 a,B138 b). The phaseerror adjusts the output of respective charge pumps (B134 a,B134 b), theoutputs of which are filtered by filters (B136 a,B136 b) and applied asthe control signals (A110,A108) to the VFOs (B130 a,B130 b). Therespective VFO outputs (A114,A106) are characterized by the equation:REF_CLK·Mx/Nx).

The output A114 of the user data synthesizer A100 is the write clockapplied to the write circuitry 9 and also the timing recovery 28 lock toreference frequency when reading user data. The output A106 of the servodata synthesizer A102 is the timing recovery 28 lock to referencefrequency when reading servo data. The synthesizer VFO control signals(A110,A108) are the coarse center frequency settings during user dataand servo data modes, respectively, for the timing recovery VFO B164 ofFIG. 7. In this manner, by matching the synthesizer VFOs (B130 a,B130 b)to the timing recovery VFO B164, the timing recovery circuit 28 operatesindependent of changes in temperature and voltage. Additionally,switching between a user and servo data synthesizer compensates for theinherently slow transient response of a single synthesizer.

Shadow Registers

The read channel gain control 50, timing recovery 28 and DC offsetcircuit 1, each comprise a filter for filtering their respective controlsignals. In the present invention, the coefficients and accumulationpaths of these filters are updated when the read channel transitionsbetween reading user data and servo data. This is necessary because theservo data is recorded at a different rate than the user data across thezones. Each filter coefficient and accumulation path has a pair ofregisters—a first register to hold the value for user data mode, and asecond register to hold a value for servo data mode. The U/S(user/servo) control signal selects the appropriate shadow registeraccording to the current mode.

The DC offset circuit 1, shown in FIG. 5, comprises a decimation filterB138 for filtering the equalized sample values 32 to compute the DCoffset in the read signal. Operation of the decimation filter B138 isdescribed in the above referenced co-pending U.S. patent applicationSer. No. 08/341,251. Basically, the DC offset is computed by adding thesample values of two consecutive opposite polarity pulses. The DC offsetis applied over line B140 to an integrating filter comprised of amultiplying coefficient B142 and an integrating accumulation path B144.When the read channel is reading user data, the U/S control signalselects a C_USER B146 a coefficient input to the multiplier B142. Whenthe read channel encounters a servo wedge, the U/S control signalselects a C_SERVO B146 b coefficient, saves the value stored in a delayregister B148 of the accumulation path B144 into an A_USER register B152a, and loads the value saved in an A_SERVO register B152 b into thedelay register B148 of the accumulation path B144. After processing theservo wedge, the filter's multiplier B142 and accumulation path B144 areupdated for reading user data in a similar manner. As a result, thefilter's frequency response is dynamically updated and the transientresponse between reading user data and servo data is minimized.

The gain control circuit 50, shown in FIG. 6, also comprises anintegrating filter having a multiplier B148 and an accumulation pathB150 which are updated between user data and servo data modes in thesame manner as the integrating filter in the DC offset circuit 1. Othergain control components shown in FIG. 6 include: a slicer B141 for fast(but less accurate) sample value estimation according to Table B2; adata gain error detector B143 for computing the gain error when readinguser and servo data; a servo address mark (SAM) gain error detector B145for computing the gain error when searching for the SAM during spin up;and a multiplexor B147 for selecting between the data and SAM gain errordetectors as controlled by the servo address mark detector A126 of FIG.3 over line A118.

Similar to DC offset 1 and gain control 50, the timing recovery circuit28, shown in FIG. 7, comprises a filter B160 for filtering the timingrecovery control signal. The timing recovery filter B160 is a PID filtershown in FIG. B that has four coefficient multiplying paths (B170 a,B170b,B170 c,B170 d) and an accumulation path B172 that are updated in thesame manner as the DC offset 1 and gain control 50 circuits. Operationof the PID filter B160 is described in the above referenced co-pendingU.S. patent application Ser. No. 08/341,251.

The timing recovery circuit 28 of FIG. 7 also comprises a zero phasestart circuit B162 for delaying operation of the VFO B164 between lockto reference and the start of acquisition until a zero crossing isdetected, thereby minimizing the initial phase error between thesampling rate and baud rate of the acquisition preamble. Phase delayshadow registers (B166 a,B166 b) compensate for delays in the zero phasestart circuit B162 and delays in the data path after the analog receivefilter 20.

The operating frequency of the timing recovery VFO B164 is adjusted by acontrol signal B161 at the output of adder B163. The control signal B161is the sum of three signals: a bias value B165 that compensates forprocess differences between the synthesizer VFOs (B130 a,B130 b) and thetiming recovery VFO B164, a fine center value B167 that is the filteredphase error output from the PID filter B160, and a coarse center value64 output from the user or servo synthesizer (A100,A102) depending onthe user/servo mode. Shadow registers (B165 a,B165 b) control theoperating range of the timing recovery VFO B164 corresponding to thechannel data rate (CDR) of the current zone.

The U/S control signal selects shadow registers (B165 a,B166 a,B168 a)when reading user data and registers (B165 b,B166 b,B168 b) when readingembedded servo data.

Other timing recovery components shown in FIG. 7 include: an expectedsample value generator B151 for generating expected samples X(n) duringacquisition mode; a slicer B141 for generating estimated samples ˜X(n)during tracking mode; a multiplexor B153 for selecting between theoutput of the expected sample generator B151 and the slicer B141; aphase error detector B155 for generating a phase error duringacquisition and tracking; a frequency error detector B157 for generatinga frequency error during lock to reference and acquisition; and amultiplexor B159 for supplying the unequalized sample values 25 duringacquisition and the equalized sample values 32 during tracking over lineB149.

The initial values for the shadow registers are determined through acalibration procedure which measures the optimum settings for each zone.When the read head passes into a new zone, the calibrated settingscorresponding to the new zone are loaded into the shadow registers.

Data/Servo Sync Detector

After acquiring the preamble (68,5) (shown in FIG. 2B), a data/servosync mark detector A120 of FIG. 3 searches for the sync mark (70,7)which demarks the beginning of the user or servo data fields. When thesync mark (70,7) is detected, the data/servo sync detector A120 enablesoperation of the RLL data decoder 36 or the RLL servo decoder A122 inorder to frame the user or servo data fields.

The data/servo sync mark detector A120 detects the sync mark (70,7) bycorrelating a target sync mark with the estimated bit sequence b(n) 33from the discrete time sequence detector. In order to minimize theprobability of early misdetection, the sync mark (70,7) is selected tohave a minimum correlation with the sync mark (70,7) concatenated withthe preamble (68,5). It is also selected for maximum probability ofcorrect detection when the sync mark is corrupted by errors due tonoise. This is accomplished with a computer search program whichsearches for an appropriate sync mark by correlating a target sync markwith shifted values of the target sync mark appended to the preamble.The search program also correlates the target sync mark with corruptedversions of the sync mark appended to the preamble.

Operation of the correlation process is understood with reference toFIG. 9. The estimated bit sequence b(n) 33 is shifted into a shiftregister C100 and the target sync mark (servo or data) is loaded intoregister C102. Registers C100 and C102 are programmable to accommodatevarious sync mark lengths. The corresponding bits of registers C100 andC102 are correlated (using an exclusive-nor gate not shown) and summedwith an adder C104. A threshold comparator C118 compares the output ofthe adder C104 to a predetermined programmable threshold and outputs athreshold correlation signal C106. The threshold correlation signal C106is enabled through an AND gate C108 by a control signal A124. The outputC114 of the AND gate C108 is applied to the RLL decoder framing signals(A121,A119) through de-multiplexor C116 according to the state of theU/S control signal. The control signal A124 for enabling the thresholdcorrelation signal C106 is understood in relation to the operation ofthe timing recovery circuit 28, an overview of which is provided in FIG.7.

In FIG. 7, the output 23 of a variable frequency oscillator (VFO) B164controls the sampling clock of a sampling device 24 which is typicallyan analog-to-digital converter (A/D) in digital read channels. Afrequency error detector B157 and phase error detector B155 control thefrequency of the VFO B164, and a loop filter B160 provides control overthe closed loop characteristics. A multiplexor B159 may select theunequalized sample values 25 during acquisition, and the equalizedsample values 32 during tracking. From the sample values received overline B149, the frequency error detector B157 generates a frequencyerror, and the phase error detector B155 generates a phase error. Thephase error is also computed from expected sample values X(n) from anexpected sample generator B151 during acquisition, and estimated samplevalues ˜X(n) from a sample value estimator B141, such as a sliceraccording to Table B2, during tracking.

Referring again to FIG. 2B, before acquiring the acquisition preamble(68,5) the phase-lock-loop first locks onto a predetermined nominalsampling frequency for the zone where the current track is located. Inthis manner, the phase-lock-loop is close to the desired acquisitionfrequency when it switches to acquisition mode. As previously mentioned,the acquisition preamble (68,5) is processed during acquisition mode inorder to lock the PLL to the desired sampling phase and frequency beforesampling the user or servo data fields (72,3). Once locked onto theacquisition preamble, the phase-lock-loop switches into tracking modeand, after detecting the sync mark (70,7), begins tracking user or servodata (72,3).

A data generator 4, connected to the input of the precoder 10, outputs aseries of “1” bits to generate a 2T training preamble sequence at theoutput of the precoder 10 of the form (1,1,0,0,1,1,0,0,1,1,0,0, . . . ).This 2T preamble maximizes the magnitude of a PR4 read channel, andduring acquisition, it is “side sampled” to generate the followingsample sequence:

(+A,−A,−A,˜A,+A,+A,−A,−A,+A,+A,−A,−A, . . . ).

FIG. 10A shows the 2T preamble “side sampled” with the expected samplesC120 in relation to the signal samples C122 and a corresponding phaseerror τ. FIG. 10B shows an implementation of the phase error detectorB155 and the expected sample value generator B151 of FIG. 7. To adjustthe initial sampling timing phase, the phase error detector B155computes a timing gradient which minimizes the mean squared errorbetween signal sample values and expected sample values. The timinggradient value Δt C124 is computed as:

Δt(n)=Y(n−1)·X(n)−Y(n)·X(n−1)

where Y(n) are the signal sample values B149 and X(n) are the expectedsample values C126.

Referring again to FIG. 10B, the outputs (C138,A124) of a 2-bit counterC128 correspond to the expected “side sampled” preamble sequence:

00→+A,−A

01→−A,−A

10→−A,+A

11→+A,+A.

The expected sample value is scaled to |A|=1 so that the multipliers(C130 a,C130 b) of the phase error detector B155 multiply by +1, −1 or0. Thus, the expected sample values X(n) C126 are two bits wide in orderto represent the ternary values:

(00=0, 01=1, and 11=−1).

A multiplexor C132, responsive to the outputs (C138,A124) of the counterC128, selects the expected sample values X(n) C126 which correspond tothe current state.

The counter C128 is loaded C134 with an initial starting state by logicC136 in response to two consecutive sample values Y(n) B149. The counteroutput bits C0 C138 and C1 A124 are initialized to:

C0=sgn(Y(n−1); and

C1=sgn(Y(n)) XOR sgn(Y(n−1))

where sgn(x) returns a 1 if x is positive and 0 if negative. Table C2shows the “side sampled” starting state values loaded into counter C128corresponding to the two consecutive sample values.

After the counter C128 is loaded with the initial starting state, itsequences through the states according to the expected samples in the 2Tpreamble at each sample clock 23. The four possible sequences are:

(+A,−A,−A,+A,+A −A, . . . )

(−A,−A,+A,+A,−A,−A, . . . )

(−A,+A,+A,−A,−A,+A, . . . )

(+A,+A,−A,−A,+A,+A, . . . )

As a result, a “hang up” problem associated with the prior art isavoided, and in addition, the state of counter C128 can beadvantageously used in the selection and detection of the sync mark(70,7).

If the 2T acquisition preamble (68,5) always ends with two positivesamples or two negative samples (e.g., samples C120 d in FIG. 10A), thenthe sync mark (70,7) will be completely loaded into register C100 ofFIG. 9 only when the counter C128 of FIG. 10B is in state (−A,−A) orstate (+A,+A) which corresponds to counter C128 outputs 01 and 11. Thatis, register C100 of FIG. 9 will contain at least one bit of theacquisition preamble preceding the sync mark if the counter is in state(+A,−A) or (−A,+A) which corresponds to counter outputs 00 and 10.Therefore, the data/servo sync mark detector A120 should be enabled onlywhen the counter C128 is in state (+A,+A) or state (−A,−A) (i.e., onlyat every other sample period). As shown in FIG. 9, the Cl counter outputA124 is applied to the AND gate C108 to enable the threshold correlationsignal C106 at every other sample period.

To ensure that the acquisition preamble (68,5) always ends in thedesired phase state (such as two positive samples or two negativesamples), the state of the precoder 10 is initialized to the appropriatevalue when writing the preamble (68,5) to the disk. For a PR4 readchannel, for example, the delay registers in the 1/1+D² precoder areinitialized to zero and an even number of 1 bits are output by the datagenerator 4 to ensure that the preamble ends in either two positivesamples or two negative samples.

Enabling the data/servo sync mark detector A120 at every other sampleperiod aids in the computer search for the optimum fault tolerant syncmark. The search program can search for minimum correlation between thesync mark and shifted versions of the sync mark concatenated with thepreamble at every other shift rather than at every shift. This increasesthe probability of finding a sync mark having a higher degree of faulttolerance.

The sync mark detection technique of the present invention can be easilyextended to search for the sync mark at every fourth sample periodrather than at every other sample period. This requires that thepreamble always ends in the same two sample values (i.e., the preambleends with the counter C128 always in one out of the four possiblestates). Further, this technique can easily be extended for use withother preamble formats (e.g., 3T, 4T, 6T, etc) and with other types ofPR read channels (e.g., EPR4 and EEPR4).

In an alternative embodiment shown in FIG. 11, the data/servo sync markdetector A120 processes two bits of the detected sequence 33 at a time.The target sync mark C102 of FIG. 9 is separated into in an even and oddinterleave and stored in an even register C150 and an odd register C152,respectively. Control logic C140 loads the even and odd interleaves(C142,C144) of the detected sequence 33 into respective shift registers(C146,C148) in response to the enable signal A124 from the counter C128.The control logic C140 delays loading the shift registers (C146,C148)with the detected sequence 33 until the counter C128 of FIG. 10B is ineither state (−A,−A) or state (+A,+A) which corresponds to counter C128outputs 01 and 11.

In yet another embodiment not shown, the data/servo sync mark detectorA120 correlates estimated sample values with expected sample values thatcorresponded to the target sync mark. For the purpose of thisdisclosure, then, the data/servo sync mark detector A120 is specified,in general, as generating channel values in response to the discretetime sample values and correlating the channel values with target valuesof a target sync mark.

The operation of the data/servo sync mark detector A120 can be describedmathematically by the following equation:

Y(k)=[t ₀ ,t ₁ , . . . t _(N−1) ]·[S _(k) , S _(k+1) , . . . ,S_(k+N−1)]^(t) ·I; where:

Y(k): output C114 of the data/servo sync mark detector A120;

k: the sample value index;

[t₀, t₁, . . . t_(N−1)]: the target values of the target sync mark;

[S_(k), S_(k+1), . . . , S_(k+N−1)]: the channel values;

N: length of the target sync mark;

I: a sample period interval that is equal to 1 when k modulo Q is amember of a set S and 0 otherwise, where Q is a predetermined integernot equal to 1.

For the 2T acquisition preamble (68,5) described in the above examplewhere the output of the data/servo sync mark detector A120 is enabled atevery other sample value, Q=2 and the set S={0}.

Asynchronous Servo Address Mark Detector

When the disk drive is initially turned on, the read head is launchedfrom a parked position, normally near the center of the disk, radiallyout over the rotating medium. The initial head location with respect tothe tracks and servo wedges is unknown. It is important to determine thelocation of the servo wedges quickly and start servoing before the headcrashes into the enclosure of the disk drive. Synchronous detection ofthe servo wedges is not possible when the locations of the servo wedgesare unknown because timing recovery cannot locate and acquire the servowedge preamble. Therefore, a special sequence of bits (normallycomprised of a long sequence of “0” bits) referred to as the servoaddress mark is recorded in at least one of the wedges. An asynchronousservo address mark detector A126 of FIG. 3 searches for this specialsequence of bits and generates a control signal A118 when found. Oncethe servo address mark has been detected, the read channel can locateand acquire the remaining servo wedges.

When searching for the servo address mark, the analog and discrete timeequalizing filters 20 and 26 are first initialized with the calibrationvalues corresponding to the highest data rate of the inner zone (whichis typically the user data rate). The channel is normally equalized tothe user data rate, rather than the servo data rate, to prevent falsedetection of the servo address mark in the user data field. The userdata synthesizer A100 is programmed to a predetermined frequency higherthan the servo data rate, and multiplexor A112 selects the centerfrequency control signal A110 from the user data synthesizer so that thetiming recovery circuit 28 samples 24 the read signal 62 at thisfrequency.

Because the servo address mark is detected asynchronously, the user andservo data gain error detector B143 of FIG. 6 cannot be used whenseeking for the servo address mark. Instead, a SAM gain error detectorB145 operates according to an algorithm that is less sensitive toamplitude fluctuations over long blocks of data. Basically, the SAMerror detector attempts to adjust the gain of the read signal accordingto a predetermined set point and the maximum absolute sample value overa programmable block length. The SAM gain error equation is:

Gain_Error=Set_Point−Max(|Sample_Value(n)|); n=k→k+N

where the Set_Point and N are programmable.

When searching for the servo address mark, a SAM detect signal A118 fromthe servo address mark detector selects the output of the SAM gain errordetector B145 as the input into the gain control filter of FIG. 6. Oncethe servo address mark has been detected, control line A118 selects, asinput to the gain control filter, the output of the data gain errordetector B143.

Servo Control

The features of the present invention interact with and respond to theservo control information contained in an embedded servo field 3 asshown FIG. 2B. Typically, the servo field 3 contains digital information(such as the track ID) for determining the inter-track position of theread head, and servo burst information for determining the centerlineoffset of the read head. The servo controller processes the inter-trackinformation when seeking to a new track and the servo bursts whentracking the centerline of a selected track during a read or writeoperation.

Referring to FIG. 12, the read head E20 generates an analog signal as itpasses over an “A” off-track burst E36, a “B” off-track burst E38, a “C”off-track burst E40 and a “D” off-track burst E42. The dibits of the B,C and D bursts E38-E42 are located at positions off of or to the side ofthe track center line E18. When the dibits of the A, B, C and D burstsE36-E42 are read by the head E20, four different analog signals result,depending on the physical position of the head E20 relative to thebursts. The analog nature of the signals derived by the bursts E36, E38,E40 and E42 are differentiated by different magnitudes.

The dibit patterns of the off-track bursts E36, E38, E40 and E42 arevery accurately positioned or written to the disk surface using a laserinterferometer, laser positioning system or other suitable technique.The dibit off-track bursts E36, E38, E40 and E42 are commonly located atpredetermined locations with respect to the track center line E18, asshown in FIG. 12. In this example, each track includes a C burst E40 anda D burst E42 positioned adjacent to but on opposite sides of the trackcenter line E18. Each track also includes either an A burst E36 or a Bburst E38. For example, each track having an even track number may havean A burst E36, while the odd numbered tracks on each side of the evennumbered track include a B burst E38. Track center lines E18 a and E18 brespectively represent the track numbers N−1 and N+1 of the track numberN represented by the center line E18. With the alternating occurrence ofthe A and B bursts E36, E38 on adjacent tracks E18, and the consistentpositional relationship of the C and D bursts E40, E42 on each trackE18, there is no overlap or conflict in the position of the bursts onthe tracks.

The derivation of the different magnitude analog off-track signals bythe head E20 reading the bursts E36, E38, E40 and E42 can be understoodby reference to FIG. 12. The head E20 is shown positioned directly abovethe track center line E18 as the off-track servo bursts E36, E38, E40and E42 approach due to the rotation of the disk. At time tA, the Aburst E36 will pass directly beneath head E20. At time tB, the B burstsE38 from the adjacent track center lines E18 a and E18 b will passsubstantially to sides of the head E20. At time tC, the C burst E40 willpass under approximately one-half of head E20, while at time tD, the Dburst E42 will also pass beneath approximately the other one-halfportion of head E20.

The dibit magnetic reversals of the bursts E36, E38, E40 and E42 inducealternating electrical signals when the head passes over the bursts. Thealternating analog signals are then typically amplified, full-waverectified, peak detected, and sampled and held in a conventional readwrite channel (not shown). Alternative techniques, such as integrationof the detected analog signals, may be utilized in the read writechannel rather than full wave rectification and peak detection.

The analog magnitude of the induced signals corresponds to the extent ofthe influence of the magnetic dibits on the head E20 as the off-trackbursts pass under the head E20. The signals derived from the bursts E36,E38, E40 and E42 are essentially related to the amount of area of thebursts which pass directly underneath the head E20. The amplitude sensedfrom the off-track bursts will be a maximum when the bursts are in themost direct alignment with head E20. For example, since head E20 is indirect alignment with the center line E18, as shown in FIG. 12 at timetA, a maximum amplitude signal will be derived by the head E20 and heldby the read write channel from the A burst E36 passing under the headE20. The relatively high amplitude analog signal results because of themaximum interaction of head E20 with the magnetic transitions of the Aburst E36 due to the center liner E18 alignment of the head E20 over theA burst E36.

Conversely, the complete off-track alignment of the B bursts E38 resultsin the head E20 sensing little or no signal from the passage of the Bbursts E38 at time tB. As head E20 remains on the center line E18, itwill derive an approximately half amplitude signal (relative to themaximum signal) at time tC from the C burst E40 and an approximatelyhalf amplitude signal at time tD from the D burst E42.

If instead of the example shown in FIG. 12, the head E20 was aligned onone of the adjacent odd numbered tracks E18 a or E18 b, little or nosignal would be derived at time tA because the head E20 would pass tothe sides of the A bursts E36, but a maximum signal would be derived attime tB because the head E20 would pass directly over the B burst E38 onthe track center line E18 a or E18 b. The signals derived at times tCfrom the C burst E40 and at time tD from the D burst E42 would beapproximately one half of the value of the maximum value, as was thesituation on the even numbered track.

The signals derived from the A, B, C and D bursts E36-E42 on each trackE18 are sometimes referred to a quadrature signals or quadratureinformation. Based on the quadrature signals and the track number, anappropriate control signal may be conveniently derived from theappropriate quadrature signals. Derivation of the control signal isbased on the largest one of the quadrature burst signals. It should alsobe recognized that different patterns and sequences of servo off-trackbursts may be employed to detect off-track information other than theabove described quadrature control system technique, but the principlesdescribed above with respect to the quadrature technique may be adaptedto other types of control systems using position defined fields toderive analog signals indicative of the position of the head E20.

With reference now to FIG. 13, a conventional analog peak detector readchannel E151 is shown incorporating an analog pulse detector E145 andservo burst area detect E146 circuits to supply signals representativeof servo data E147 and servo burst amplitudes E148, respectively, to anassociated servo controller E152 incorporating an on-board ADC E149. Thehead signal appearing on line E141 is supplied to the analog pulsedetector E145 for detecting the user data E100 and the servo data E147.The head signal E141 is also supplied to the analog area detect or peakdetect E146 circuit for measuring the servo burst amplitudes E148.Digital servo data on line E147 is passed directly to the servo controldigital processor E150 from the analog pulse detector E145 while ananalog signal representative of the burst amplitudes on line E148 ispassed to the ADC E149 of the servo controller E152.

Referring now to FIG. 14, a sampled amplitude read channel E161 andassociated servo controller E162 in accordance with the presentinvention is shown using discrete time servo demodulation which is moreefficient for sampled amplitude read channels and further reduces theduplication of the corresponding portions of the ICs illustrated in FIG.13. The sampled amplitude read channel E161 operates in the same manneras described in FIG. 1. In addition, a multiplexor E214 selects eitherthe ADO output 25 or the discrete time equalizer output 32 as the inputto a discrete time asynchronous area detect circuit E206 and anasynchronous discrete time pulse detector E200. A multiplexor E220selects as the detected servo data E147 sent to the servo controllerdiscrete time processor E208 either the output of the asynchronous pulsedetector E202 or the detected data {circumflex over ( )}b(n) 33 from thesynchronous discrete time sequence detector 34.

As can be observed in FIG. 14, the ADC E149 of the conventional servocontroller E152 in FIG. 13 is obviated by the present invention becausethe servo burst information is sampled using the read channel ADC 24 ofFIG. 14 and processed in discrete time. Consequently, the interfacebetween the sampled amplitude read channel E161 and servo controllerE162 is all digital. Another illustrated advantage is to detect theservo data E147 using the discrete time pulse detector E200 alreadyprovided in the sampled amplitude read channel E161 for timing and gaincontrol. Thus, the analog pulse detection circuitry in conventionalservo demodulation is also obviated. The only additional circuitrynecessary to implement the servo demodulation technique of the presentinvention is a discrete time area detect circuit E206 which providesdiscrete servo burst amplitude information to the servo controller E162over line E104.

Digital Servo Data Detection

With reference now to FIG. 15, shown is an asynchronous pulse detectorin accordance with the present invention for use in conjunction with thedigital servo demodulator circuit. The samples Xn are supplied to thepositive inputs of comparators E301 and E305 as well as to the negativeinput of comparator E302. A delayed sample is supplied to the negativeinput of comparator E305 as shown. The outputs of comparators E301,E302, and E305 are supplied to a logic block E306 to provide the signalsSBITPn and SBITn which correspond, respectively, to the presence andpolarity of detected pulses.

The servo signal is sampled by the ADC 24 of the read channel E161(shown in FIG. 14) at a rate faster (typically more than 4 times faster)than the rate of pulses in the servo data fields. ADC 24 samples Xn arecompared to thresholds T+ E303 and T− E304 to create the bits Hn and Ln,respectively. If the sample Xn is greater than T+ E303 then Hn isactive. If, on the other hand, Xn is less than T− E304 then Ln isactive. In addition, samples Xn are compared to the previous samplesXn−1 to create the bits En and Dn. If Xn is greater than Xn−1 then Dn isactive. Alternatively, if Xn is equal to Xn−1 then En is active.

The logic block E306 illustrated in FIG. 15 is shown in more detail inFIG. 16. The servo bit detector can operate in four modes. Mode 1: Peakdetection with polarity qualification; Mode 2: Peak detection withoutpolarity qualification; Mode 3: Threshold detection with polarityqualification; and Mode 4: Threshold detection without polarityqualification. The output data bits SBITPn and SBITn are derived by thecircuit of FIG. 16.

The signal En is delayed through a delay element and applied to aninverting input of AND gate E405. Similarly, the signal Dn is delayedthrough a delay element and applied to a non-inverting input of AND gateE404 and an inverting input of AND gate E405. The signal Dn is alsodirectly supplied to an inverting input of AND gate E404 and anon-inverting input of AND gate E405.

The output of AND gate E404 is supplied as one input to OR gate E406which has its other input coupled to a Threshold/˜Peak signal E420.Threshold/˜Peak signal E420 is also provided as an input to OR gateE407. The remaining input to OR gate E407 is taken at the output of ANDgate E405. A ˜Polarity signal E422 is supplied as one input to OR gateE408 a and OR gate E408 b. The outputs of OR gates E406 and E408 a inconjunction with the delayed threshold signal Hn−1 are supplied asinputs to three input AND gate E410 which has its output coupled as oneinput to OR gate E412. In like manner, the outputs of OR gates E407 andE408 b in conjunction with the delayed threshold signal Ln−1 aresupplied as inputs to AND gate E411 having its output comprising theremaining input to OR gate E412.

The signal Hn is delayed by one clock and then supplied as the firstinput to multiplexer (“mux”) E413. The output of mux E413 is selected bythe output of OR gate E412 SBITn. The output of mux E413 is coupled to adelay register E403 as shown with the signal SBITPn derivedtherebetween. The output of the delay register E403 is provided as thesecond input to mux E413. The signal SBITPn qualifies the outputs of ORgates E408 a and E408 b such that only peaks alternating in polarity aredetected when in polarity qualification mode.

In operation, the output of AND gates E410 and E411 indicate thepresence of a positive or negative pulse. The outputs of these gates areORed E412 to generate the SBITn signal indicating the presence of apulse. In peak mode, positive and negative peaks are detected by ANDgates E404 and E405 respectively. The logic equation for positive peakdetecting AND gate E404 is:

+Peak=!Dn AND Dn−1; or,

+Peak=!(Xn>Xn−1) AND (Xn−1>Xn−2).

The logic equation for negative peak detecting AND gate E405 is:

−Peak=!En−1 AND!Dn−1 AND (Dn OR En); or,

−Peak=!(Xn−1==Xn−2) AND!(Xn−1>Xn−2) AND!(Xn<Xn−1); or,

 −Peak=(Xn−1<Xn−2) AND!(Xn<Xn−1).

From these equations, it is understood that a peak is only detected ifthere is a change of slope in the analog signal. This is illustrated inFIG. 17.

If Threshold/˜Peak signal E420 is low (peak mode), then the peak signalsfrom AND gates E404 and E405 are passed by OR gates E406 and E407. Thepeaks are qualified by thresholds Hn−1 and Ln−1 through pulse detectingAND gates E410 and E411. That is, a pulse will only be detected if Xn−1exceeds the positive or negative threshold for a positive or negativepeak, respectively, as shown in FIG. 17.

If Threshold/-Peak signal E420 is high (threshold detect mode), theoutputs of OR gates E406 and E407 are always active, and the output ofpulse detecting AND gates E410 and E411 are responsive only to thethreshold signals Hn−1 and Ln−1 and the polarity qualification signalsfrom OR gates E408 a and E408 b.

If the ˜Polarity signal E422 is low (polarity qualify mode), OR gatesE408 a and E408 b pass the polarity signal SBITPn to AND gates E410 andE411 which will then detect pulses of alternating polarity only. If the˜Polarity signal E422 is high, the outputs of OR gates E408 a and E408 bare always active thereby disabling the polarity qualification mode.

The polarity qualification signal SBITPn is generated as follows. When apulse is detected, the SBITn signal is active and selects as output ofmux E413 (which is also SBITPn) the delayed threshold signal Hn−1. Ifthe currently detected pulse is positive, then Hn−1 (and SBITPn) is highand the expected polarity of the next pulse negative. Otherwise, Hn−1(and SBITPn) is low and the expected polarity of the next pulsepositive. Delay register E403 stores the updated value of SBITPn untilthe next pulse is detected. The pulse detection AND gates E410 and D411are enabled according to the expected polarity of the next pulse throughOR gates E408 a and E408 b. That is, a positive or negative pulse willonly be detected if the polarity of signal SBITPn is opposite inpolarity of the pulse detected.

By oversampling (sampling faster than the baud rate), the detection ofservo data is relatively insensitive to the phase and frequencies of theADC 24 clock of the read channel E161 shown in FIG. 14. The flexibilityof the detector modes provides good performance over a variety ofconditions including servo data rates and filtering. Alternatively, thesequence detector 34 (commonly a Viterbi type sequence detector) of thesampled amplitude read channel E161 may be used to detect servo datasynchronous to the baud rate in the same way user data is detectedrather than using a pulse detector. Detecting the servo data with asequence detector can be more accurate and more efficient, especially ifthe sampled amplitude read channel does not otherwise use a pulsedetector E200.

Digital Servo Burst Measurement

With reference now to FIG. 20, an asynchronous servo burst demodulatorfor use in conjunction with the present invention is shown. Theasynchronous servo burst demodulator comprises a nonlinearity circuitE701 for performing an absolute value (ABS) or interpolate functioncoupled to an accumulator E702, which in turn is coupled to a number ofregisters E703 designated A, B, C and D. In burst amplitude detection,the ADC 24 (shown in FIG. 14) signal samples are first passed throughthe nonlinearity circuit E701 and then summed in the accumulator E702.The resulting amplitude measurements are held in the register E703depending on the burst selection signal Bsel.

The accumulation of samples tends to cancel errors due to ADCquantization. This quantization error is reduced with the number ofsignal samples accumulated leaving a residual error which appearssimilar to a small amount of added noise. However, this accumulatederror is much less than the typical noise otherwise experienced in thedata channel. The digital area detection technique of the presentinvention, therefore, has advantages over an alternative digital burstamplitude measurements with respect to noise immunity. Since noise inthe signal samples is accumulated, and effectively averaged, the noisetends to cancel as in a low bandwidth filter. Thus, for typically strongsignals, the noise performance of digital area detection is close tothat of the theoretically optimum detector. Also, accumulating the burstamplitude signal samples increases the effective resolution of thechannel ADC in proportion to the number of samples accumulated. In thismanner, the resolution of the channel ADC (typically 6 bits) iseffectively increased by 2 to 4 bits or more.

Still another advantage of a digital area detection technique is itsrelative insensitivity to direct current (“DC”) offsets in a magneticchannel servo signal. Since the signal slope through the zero level isfairly high, a DC offset in the signals tends to add about as much areato half of the pulses as it subtracts from the other half giving a netzero change in measured amplitude to a first order approximation. Yetanother advantage of a digital area detection technique is that it tendsto de-emphasize the anomalous amplitude of one pulse by averaging manypulses in a burst. This is significantly better than a peak detectiontechnique which can respond fully to a single anomalous pulse.

With reference now to FIGS. 19A-19D, the functionality of theasynchronous servo demodulator circuit of the present invention isshown. In a basic area detection function, the ADC 24 (shown in FIG. 14)signal samples are full wave rectified by an absolute value operationand then accumulated. As shown in FIG. 19B, the rectified signal E602has steep notches at zero crossing E603 of the servo burst E601 shown inFIG. 19A. With some sample rates, many signal samples can fall at thebottom of these notches or just outside of the notches givingsignificantly different burst amplitude measurements depending on thephase of the servo burst relative to the ADC 24 clock.

The asynchronous servo demodulator circuit of the present inventionincorporates a predetermined subset of five techniques overcoming thisproblem of sampling phase sensitivity: (a) control the samplingfrequency relative to the burst frequency, (b) clock dither andfrequency control, (c) squaring, (d) interpolation (e) and control ofthe accumulation window. Each technique tends to reduce the variation ofthe burst amplitude measurement with phase.

The simplest technique for reducing phase sensitivity is to control thesample frequency relative to the servo burst frequency so that only afew samples can line up with the zero crossings in a single servo burst.Nevertheless, this may not be practical in some disk drives since thereare often limitations in the burst rates which may be written. Moreover,the servo burst frequency varies during read back with the angularvelocity of the disk relative to the read/write head.

With reference additionally to FIG. 18, another technique which reducesphase sensitivity is to “dither” or “sweep” the ADC sampling frequencyso that few samples can line up with the zero crossings in one servoburst. Utilizing this technique, the ADC 24 sampling frequency ischanged over some set of frequencies within a small fraction of anominal clock rate. If one of these frequencies allow samples to alignwith zero crossings, it only lasts for a short time and the number ofunreliable samples is therefor minimized. Further, clock control tendsto randomize the errors due to ADC quantization. This enhances theresolution improving effect of accumulating several sample values.

Dithering or sweeping the ADC 24 sampling frequency may be implementedas shown in FIG. 18 wherein a four-bit counter circuit E601 sequencesthrough 16 binary states which are passed to the frequency control busE603. This frequency control bus offsets the ADC 24 sampling frequencyby a fractional amount based on a binary number representation.

Referring specifically to FIG. 19C, another technique for reducing phasesensitivity is to use a squaring operation for rectifying the signal.Since the filtered servo burst is very near sinusoidal, after squaring,the signal E604 has the form of a sinusoid shifted so that its minimumvalue E605 is zero. Then, since the squared servo burst is accumulatedfor an integer number of cycles, the sinusoidal variation cancels (tothe first order) leaving the average value which is proportional to theservo burst energy independent of the phase. One disadvantage of thistechnique is that it requires more data precision, more complicatedcircuitry for rectification, and may require a square root operation tobe performed on the result. A further disadvantage of a squaringoperation is that it can inadvertently emphasize noise.

Referring now to the logic block diagram of FIG. 21, another techniquefor reducing phase sensitivity is interpolation. In this technique,consecutive samples are averaged to estimate the servo burst signal atthe time between signal samples. Then all samples and interpolatedsamples are rectified an accumulated to produce a burst amplitudemeasurement.

As shown in FIG. 21, an ABS ( ) or interpolate circuit E701 is shown.Circuit E701 comprises a pair of AND gates E802 a and E802 b to receivepairs of consecutive input samples Xn+1 and Xn as well as an interpolateenable signal on line E801. Processing two samples at a time allows forhalf-rate sampling. The output of AND gate E802 a is summed with theoutput of AND gate E802 b in adder E805 and divided by two with a simpleshifter E807. The output of shifter E807 is supplied to an abs ( )function logic block E809 for application to adder E811. Adder E811takes the sample Xn+1 through the absolute value operator function blockE803 and provides it as one input to adder E813.

In like manner, the output of AND gate E802 b is summed with the delayedoutput of AND gate E802 a in adder E806 for division by two throughshifter E808 for application to abs ( ) function logic block E810 andsubsequent application to adder E812. Adder E812 has as an additionalinput the signal Xn as received through the absolute value operatorfunction block E804. The output of adder E812 provides the second inputto adder E813 to produce the area signal “A.”

As shown in FIG. 21, signal samples are processed in parallel. When the“interpolate” signal on line E801 is inactive, the AND gates E802 a andE802 b block the input signals and the output of the inner “abs”operators E809 and E810 are zero. Then, the signal samples Xn+1 and Xnare rectified by the absolute value operators E803 and E804 and summedby the adders E811 and E812. This result is accumulated for a fixed evennumber of clock cycles. When the “interpolate” signal on line E801 isactive, the AND gates E802 a and E802 b pass the signal samples to theinterpolator.

In the upper branch, the signal samples Xn+1 and Xn are summed by theadder E805 and divided by two with a simple shifter E807. The resultinginterpolated sample is rectified at block E809 and added to thenon-interpolated signal in adder E811. Likewise, in the lower branch,signal sample Xn is added to the delayed signal Xn+1 (that is, Xn−1) inadder E806 and is divided by two in the shifter E808. The resultinginterpolated sample is rectified in block E810 and added to thenon-interpolated sample Xn in adder E812. The two sums are then added inadder E813 giving an output which is the sum of two rectified samplesand two rectified interpolated samples.

The results of interpolating before rectification is that more samplesare accumulated near the servo burst zero crossings filling in thenotches shown in FIG. 19D. This is close to the same result as if thesignal were sampled at twice the rate. Since more samples appear in thenotches of the rectified servo burst, the signal is more accuratelyrepresented and the resulting servo burst amplitude varies less withsampling phase. This technique has the advantage of simpler circuitrythan squaring along with better noise performance. Therefore, thepreferred implementation of digital area detection uses interpolation toreduce phase sensitivity along with frequency control and dither. Thiscombination of techniques is relatively simple to implement inaccordance with the present invention and provides robust performance.

With reference additionally now to FIG. 22, another effect which causesvariation of the servo burst amplitude measurement with sampling phaseis the alignment of the accumulation window with the servo burst signal.If the accumulation does not include exactly an integer number of pulses(that is, half-cycles), then the measurement will change depending onthe exact alignment or phase. This problem may be mitigated bytriggering the beginning of accumulation by the signal “BWIN” after theservo burst signal has had time to reach steady state and appearsinusoidal. Then, the servo burst amplitude circuit accumulates for apredetermined period of time which is very close to an integer number ofservo pulse periods.

Referring now to FIG. 22, the digital area detect circuit accumulatesrectified samples in accumulator E901 immediately after the signal“BWIN” is made active. Also, counter E902 begins counting accumulationclocks at this time, starting at zero. Then, accumulation continuesuntil the counter E902 reaches a programmed count number. At this time,the resulting accumulation is stored in one of the registers E903,depending on the value of the signal “BSEL”. This process repeats untilall servo bursts in the present field are measured. The servo burstamplitude measurements may then be read by the servo controller E162(shown in FIG. 14) from the registers E903. This may be accomplishedwith a parallel or serial digital interface (not shown).

Many changes in form and detail could be made to the present inventionwithout departing from the essential function the particular embodimentsdisclosed herein are not intended to be limiting. The scope of theinvention is properly construed from the following claims.

TABLE 1 Channel Transfer Function Dipulse Response PR4 (1 − D) (1 + D)0, 1, 0, −1, 0, 0, 0, . . . EPR4 (1 − D) (1 + D)² 0, 1, 1, −1, −1, 0, 0,. . . EEPR4 (1 − D) (1 + D)² 0, 1, 2, 0, −2, −1, 0, . . .

TABLE B2 Sample Value Slicer Output y >= T1 +1 −T2 <= y < T1 0 y < −T2−1

TABLE C2 State Y (n − 1) Y (n) C0, C1 +A, −A +y +y 00 −A, −A +y −y 01−A, +A −y −y 10 +A, +A −y +y 11

We claim:
 1. A sampled amplitude read channel for reading data recordedon a magnetic disk medium by detecting digital data from a sequence ofdiscrete time sample values generated by sampling pulses in an analogread signal from a magnetic read head positioned over the magnetic diskmedium, the magnetic disk medium comprising a plurality of concentricdata tracks wherein a data track comprises a plurality of sectors, thesampled amplitude read channel comprising: (a) a discrete time timingrecovery circuit for synchronizing the discrete-time sample values to abaud rate of the data recorded on the magnetic disk medium; (b) adiscrete time gain control circuit for adjusting an amplitude of theanalog read signal relative to a desired partial response; and (c) adiscrete time sequence detector for detecting an estimated data sequencefrom the sequence of discrete time sample values; wherein the timingrecovery circuit and the gain control circuit are reconfigured beforethe discrete time sequence detector finishes processing the discretetime sample values of a current sector so that the read channel canbegin acquiring an acquisition preamble of a next sector, therebyreducing a physical gap between sectors on the magnetic disk medium. 2.The sampled amplitude read channel as recited in claim 1, wherein: (a)the digital data comprises user data and embedded servo data; (b) theconcentric data tracks comprise user data and servo data sectors; and(c) the sampled amplitude read channel transitions between a user dataand servo data mode.
 3. The sampled amplitude read channel as recited inclaim 1, further comprising a discrete time equalizer for equalizing thediscrete time sample values according to the desired partial response,wherein the timing recovery circuit and the gain control circuit arereconfigured before the discrete time equalizer finishes processing thediscrete time sample values of a current sector so that the read channelcan begin acquiring an acquisition preamble of a next sector, therebyreducing a physical gap between sectors on the magnetic disk medium. 4.The sampled amplitude read channel as recited in claim 1, wherein: (a)the timing recovery circuit comprises a loop filter comprising aplurality of coefficients; and (b) the coefficients of the loop filterare reconfigured before the discrete time sequence detector finishesprocessing the discrete time sample values of a current sector so thatthe read channel can begin acquiring an acquisition preamble of a nextsector, thereby reducing a physical gap between sectors on the magneticdisk medium.
 5. The sampled amplitude read channel as recited in claim1, wherein: (a) the gain control circuit comprises a loop filtercomprising a plurality of coefficients; and (b) the coefficients of theloop filter are reconfigured before the discrete time sequence detectorfinishes processing the discrete time sample values of a current sectorso that the read channel can begin acquiring an acquisition preamble ofa next sector, thereby reducing a physical gap between sectors on themagnetic disk medium.
 6. The sampled amplitude read channel as recitedin claim 1, further comprising a discrete time DC offset circuit forattenuating a DC offset in the analog read signal, wherein the DC offsetcircuit is reconfigured before the discrete time sequence detectorfinishes processing the discrete time sample values of a current sectorso that the read channel can begin acquiring an acquisition preamble ofa next sector, thereby reducing a physical gap between sectors on themagnetic disk medium.
 7. The sampled amplitude read channel as recitedin claim 6, wherein: (a) the DC offset circuit comprises a loop filtercomprising a plurality of coefficients; and (b) the coefficients of theloop filter are reconfigured before the discrete time sequence detectorfinishes processing the discrete time sample values of a current sectorso that the read channel can begin acquiring an acquisition preamble ofa next sector, thereby reducing a physical gap between sectors on themagnetic disk medium.
 8. A method of reading data recorded on a magneticdisk medium by detecting digital data from a sequence of discrete timesample values generated by sampling pulses in an analog read signal froma magnetic read head positioned over the magnetic disk medium, themagnetic disk medium comprising a plurality of concentric data trackswherein a data track comprises a plurality of sectors, the method ofreading data comprising the steps of: (a) synchronizing thediscrete-time sample values to a baud rate of the data recorded on themagnetic disk medium using a discrete time timing recovery circuit; (b)adjusting an amplitude of the analog read signal relative to a desiredpartial response using a discrete time gain control circuit; (c)detecting an estimated data sequence from the sequence of discrete timesample values using a discrete time sequence detector; and (d)reconfiguring the timing recovery circuit and the gain control circuitbefore the sequence detector finishes processing the discrete timesample values of a current sector so that the read channel can beginacquiring an acquisition preamble of a next sector, thereby reducing aphysical gap between sectors on the magnetic disk medium.
 9. The methodof reading data from a magnetic disk medium as recited in claim 8,wherein the concentric data tracks comprise user data and servo datasectors, further comprising the step of transitioning between a userdata and servo data mode.
 10. The method of data from a magnetic diskmedium as recited in claim 8, further comprising the step of equalizingthe discrete time sample values according to the desired partialresponse using a discrete time equalizer, wherein the timing recoverycircuit and the gain control circuit are reconfigured before thediscrete time equalizer finishes processing the discrete time samplevalues of a current sector so that the read channel can begin acquiringan acquisition preamble of a next sector, thereby reducing a physicalgap between sectors on the magnetic disk medium.
 11. The method ofreading data from a magnetic disk medium as recited in claim 8, wherein:(a) the timing recovery circuit comprises a loop filter comprising aplurality of coefficients; and (b) the coefficients of the loop filterare reconfigured before the discrete time sequence detector finishesprocessing the discrete time sample values of a current sector so thatthe read channel can begin acquiring an acquisition preamble of a nextsector, thereby reducing a physical gap between sectors on the magneticdisk medium.
 12. The method of reading data from a magnetic disk mediumas recited in claim 8, wherein: (a) the gain control circuit comprises aloop filter comprising a plurality of coefficients; and (b) thecoefficients of the loop filter are reconfigured before the discretetime sequence detector finishes processing the discrete time samplevalues of a current sector so that the read channel can begin acquiringan acquisition preamble of a next sector, thereby reducing a physicalgap between sectors on the magnetic disk medium.
 13. The method ofreading data from a magnetic disk medium as recited in claim 8, furthercomprising a discrete time DC offset circuit for attenuating a DC offsetin the analog read signal, wherein the DC offset circuit is reconfiguredbefore the discrete time sequence detector finishes processing thediscrete time sample values of a current sector so that the read channelcan begin acquiring an acquisition preamble of a next sector, therebyreducing a physical gap between sectors on the magnetic disk medium. 14.The method of reading data from a magnetic disk medium as recited inclaim 13, wherein: (a) the DC offset circuit comprises a loop filtercomprising a plurality of coefficients; and (b) the coefficients of theloop filter are reconfigured before the discrete time sequence detectorfinishes processing the discrete time sample values of a current sectorso that the read channel can begin acquiring an acquisition preamble ofa next sector, thereby reducing a physical gap between sectors on themagnetic disk medium.